Current measurement circuit

ABSTRACT

A current measurement circuit converts a current signal I IN  into a voltage signal V OUT . The current signal I IN  is transmitted via a signal line. A shield line is arranged in the vicinity of at least a part of the signal line. A non-inverting amplifier includes an operational amplifier, and the current signal I IN  is input to its non-inverting input terminal. The output signal of the non-inverting amplifier is input to its inverting input terminal as a feedback signal. An inverting amplifier amplifies the output signal of the non-inverting amplifier with inversion so as to generate a voltage signal V OUT . An impedance circuit includes a feedback resistor R F  between the output terminal of the inverting amplifier and the non-inverting input terminal of the operational amplifier. A guard amplifier receives the electric potential at the inverting input terminal of the operational amplifier, and applies the electric potential to the shield line.

CROSS-REFERENCE TO RELATED APPLICATIONS

The present invention claims priority under 35 U.S.C. §119 to JapaneseApplication No. 2014-242019, filed Nov. 28, 2014, the entire content ofwhich is incorporated herein by reference.

BACKGROUND OF THE INVENTION

Field of the Invention

The present invention relates to a current measurement circuit.

Description of the Related Art

In order to analyze the base sequence of DNA (deoxyribonucleic acid),RNA (ribonucleic acid), or the like, a base sequence analyzing apparatus(sequencer) is employed. As a next-generation (fourth-generation)sequencer, various kinds of techniques have been sought by researchinstitutions and industries. As one of such prospective techniques, thegating nanopore sequencing technique has attracted attention.

With the gating nanopore sequencing technique, DNA or RNA is moved suchthat it passes through a gap between a pair of nanometer-orderelectrodes (nano-electrodes). The tunnel current that flows through theelectrode gap changes according to the base type (A, G, T, C) thatpasses through the electrode gap. The base sequence is determined basedon the change in the tunnel current. This technique is anticipated tohave the potential to provide a very low-cost and very compact-sizeapparatus that is capable of analyzing a base sequence. It should benoted that, in the present specification, examples of such a“nano-electrode” include sub-micro electrodes and micro electrodeshaving a larger size.

Also, as a method using a tunnel current in the same way as with thegating nanopore sequencing technique, the MCBJ (MechanicallyControllable Break Junction) method has been developed. With the MCBJmethod, a nano-electrode is formed by breaking a metal wire.

As an important element technology, such a sequencer requires a currentmeasurement device that is capable of measuring a tunnel current thatflows through a nano-electrode gap with sufficiently high precision.That is to say, such a tunnel current has a current value on the orderof several tens of pA. Accordingly, in order to judge the base type,there is a need to detect a difference in conductance on the order ofseveral ps (picosecond).

The present inventors have investigated an arrangement employing atransimpedance amplifier as a microscopic current measurement device.FIG. 1 is a circuit diagram showing a current measurement circuit 900including a transimpedance amplifier 800. The transimpedance amplifier800 includes an operational amplifier 802 and a resistor R_(F) arrangedbetween the inverting input terminal (−) and the output terminal of theoperational amplifier 802. A predetermined electric potential V_(REF)(e.g., ground voltage) is input to the non-inverting input terminal (+)of the operational amplifier 802. A capacitor C_(F) is connected inparallel with the resistor R_(F), in order to provide the circuit withstable operation.

A DUT (device under test) 810 includes a sample such as DNA, RNA, or thelike (which will collectively be referred to as “DNA” hereafter), and achip configured to house the sample. A nanochannel, nanopillarstructure, and an electrode pair are formed in the chip such that a DNAmolecule separated from the sample passes through such components. Acable 820 connects the DUT 810 and the transimpedance amplifier 800.

FIG. 2 is an equivalent circuit diagram showing an equivalent circuit ofthe current measurement circuit 900 shown in FIG. 1. The DUT 810 can bemodeled as a circuit comprising a current source 812 that generates atunnel current i_(DUT), a parasitic parallel resistor R_(DUT), and aparasitic parallel capacitor C_(DUT).

The cable 820 includes a first line 822 that connects one end 814 of theDUT 810 and the inverting input terminal of the operational amplifier802, and a second line 824 that connects the other end 816 of the DUT810 and the non-inverting input terminal of the operational amplifier802. Here, C_(CAB) represents a parasitic capacitance that occursbetween the two lines 822 and 824. In a case in which the cable 820 isconfigured as a coaxial cable, a parasitic capacitance of 10 pF occursin increments of 10 cm of the cable 820.

Various kinds of parasitic capacitances occur in the input stage of thetransimpedance amplifier 800. For example, C_(PRO) represents aparasitic capacitance that occurs due to an ESD protection element 830such as a diode, ESD suppresser, or the like. The operational amplifier802 is represented by an equivalent circuit comprising an idealamplifier 804 and various kinds of parasitic capacitances. Here, CMN andCMP each represent a common input capacitance, and CD represents adifferential input capacitance. It should be noted that, in FIG. 2, theresistance values and capacitance values are shown for exemplarypurposes only.

The DC transimpedance of the transimpedance amplifier 800 is representedby the following Expression.20×log₁₀(R _(F))(dB)  (1)

For example, in a case in which R_(F)=1 GΩ, the transimpedance amplifier800 has a DC transimpedance of 180 dB.

Such a DNA sequencer is required to identify the kinds of bases withrespect to an enormous number of base pairs, the number of which is onthe order of several billion. A fourth-generation DNA sequencer isrequired to provide a measurement time on the order of 1 ms per base.However, it is difficult for such a fourth-generation DNA sequencer toidentify a base based on a single measurement due to the influence ofnoise. Thus, the tunnel current is measured multiple times during themeasurement time of 1 ms, and the base is identified using a statisticalmethod. Specifically, the base is identified based on a histogram of themeasurement results, for example. For example, in a case in which thetunnel current is measured 100 times during the measurement time of 1ms, such an arrangement requires a sampling rate of 100 ksps. In thiscase, the transimpedance amplifier is required to have a bandwidth ofseveral hundreds of kHz to several MHz, which is estimated givingconsideration to a margin.

Here, examination will be made regarding the frequency characteristicsof the transimpedance amplifier 800 shown in FIG. 2. The cutofffrequency f2 is represented by the following Expression (2).f2=1/{2πR _(F)×(C _(F) +C _(S) /A _(OL))}  (2)

It should be noted that C_(S) is represented byC_(S)=C_(DUT)+C_(CAB)+C_(PRO)+CD. Here, A_(OL) represents the open loopgain of the operational amplifier. As can be understood from Expression(2), in order to raise the cutoff frequency f2, an approach can beemployed in which C_(F) and C_(S) are each reduced, and A_(OL) is raisedover a wide bandwidth. Here, C_(S) will be referred to as “input shuntcapacitance”. In a case in which the open loop gain A_(OL) issufficiently large, and the input shunt capacitance C_(S) issufficiently small, Expression (2) is approximated by the followingExpression (3).f2≈1/{2πR _(F) ×C _(F)}  (3)

For example, in a case in which R_(F)=1 GΩ, and C_(F)=10 fF, f2=15.9 kHzis obtained based on Expression (3).

However, the tunnel current has a very small current value. Thus, such atunnel current is affected by measurement system noise, which is aproblem. FIG. 3 is a diagram showing the noise characteristics of thetransimpedance amplifier. In order to detect such a microscopic current,such an arrangement requires a resistor R_(F) on the order of severaltens of MΩ to several TΩ. Accordingly, in the low-frequency range,thermal noise due to the resistor R_(F) becomes dominant.V _(NOISE)=√(4×k×T×R _(F))

Here, T represents the temperature, and k represents the Boltzmannconstant. This expression represents the voltage noise density per unitfrequency.

In a case in which R_(F)=1 GΩ, and T=27 degrees, V_(NOISE)=4.1 μV/√Hz(which is also represented by “V/rtHz”) is obtained.

The transimpedance amplifier 800 imposes a band limit on the thermalnoise with the aforementioned cutoff frequency f2 as the boundary. Thus,in the high-frequency range that is higher than the cutoff frequency f2,the noise from the transimpedance amplifier 800 becomes dominant ascompared with the thermal noise that occurs in the resistor R_(F). Inthe high-frequency range, the noise gain of the amplifier isproportional to (C_(F)+C_(S)+CM+CD)/C_(F). Thus, in order to reduce thenoise, C_(S), CM, and CD must be reduced, and C_(F) must be raised.However, an increase in C_(F) leads to a reduction in the cutofffrequency f2, which is opposite to a requirement of increasing thebandwidth. Thus, there is a need to design the capacitor C_(F) to haveas small a value as possible in a range so as to ensure systemstability. As described above, with the transimpedance amplifier shownin FIG. 1, there is a tradeoff relation between the bandwidth (cutofffrequency) to be raised and the noise to be reduced. That is to say, itis difficult to provide both a wide bandwidth and low noise.

In particular, the operational amplifier has a very high inputimpedance. Accordingly, the transimpedance amplifier is greatly affectedby electric-field noise. In order to reduce such noise, a technique isknown in which the signal line is covered by a shield, and the electricpotential at the shield is controlled. However, in a case in which sucha shield is provided as an additional component to the transimpedanceamplifier which is required to provide a high-speed operation, thisleads to an increase in the parasitic capacitance due to the signalline. In addition, an amplifier that drives the shield involves acapacitance. Thus, such an arrangement leads to a narrow bandwidth and areduced operation speed.

SUMMARY OF THE INVENTION

The present invention has been made in order to solve such a problem.Accordingly, it is an exemplary purpose of an embodiment of the presentinvention to provide a current measurement circuit which providesreduced noise and/or a widened bandwidth.

An embodiment of the present invention relates to a current measurementcircuit that converts a current signal into a voltage signal. Thecurrent measurement circuit comprises: a signal line via which thecurrent signal is transmitted; a shield line arranged in the vicinity ofat least a part of the signal line; a non-inverting amplifier comprisingan operational amplifier having its non-inverting input terminalsupplied with the current signal the signal line and its inverting inputterminal supplied with its output signal as a feedback signal; aninverting amplifier that amplifies the output signal of thenon-inverting amplifier with inversion so as to generate the voltagesignal; an impedance circuit comprising a feedback resistor arrangedbetween an output terminal of the inverting amplifier and thenon-inverting input terminal of the operational amplifier; and a guardamplifier that receives an electric potential at the inverting inputterminal of the operational amplifier, and that applies the electricpotential to the shield line.

By providing a two-stage configuration comprising the non-invertingamplifier and the inverting amplifier, such an arrangement is capable ofreducing the effect of the differential input capacitance that occurs inthe operational amplifier included in the non-inverting amplifierconfigured as a first-stage amplifier. Furthermore, by providing theshield line in the vicinity of the signal line, such an arrangement iscapable of reducing the capacitance that occurs between the signal lineand the ground. With such an arrangement, by virtually grounding theoperational amplifier, the electric potential at the inverting inputterminal of the operational amplifier becomes substantially the same asthat at the non-inverting input terminal thereof. The signal line isconnected to the non-inverting input terminal of the operationalamplifier. Furthermore, the electric potential at the inverting inputterminal of the operational amplifier is applied to the shield line viathe guard amplifier. Thus, the electric potential at the signal linebecomes substantially the same as that at the shield line. This allowsthe effects of the capacitance that occurs between the signal line andthe shield line to be reduced. Furthermore, the input of the guardamplifier is connected to the inverting input terminal of theoperational amplifier included in the non-inverting amplifier. Thus, theinput capacitance that occurs in the guard amplifier is not coupled withthe signal line. Thus, such an arrangement does not lead to an increasein the capacitance as viewed from the signal line. That is to say, itcan be said that the effect of the guard amplifier on the cutofffrequency is substantially zero.

Thus, such an embodiment is capable of greatly reducing the effects ofthe parasitic capacitance as viewed from the signal line, therebyreducing the noise gain. Furthermore, by providing such a two-stageamplifier configuration, such an arrangement provides a high gain over awide bandwidth while suppressing an increase in noise.

Also, the current measurement circuit may further comprise a correctionamplifier that corrects frequency characteristics of an output signal ofthe inverting amplifier.

As described above, by providing such a two-stage configurationcomprising the non-inverting amplifier and the inverting amplifier, suchan arrangement is capable of providing reduced noise gain in ahigh-frequency range. Thus, by providing the correction amplifier thatboosts such a high-frequency signal component, such an arrangementprovides a high signal gain over a wide bandwidth while maintaining thenoise gain at the same level as that provided by conventionalarrangements.

Also, the current measurement circuit may further comprise a shield thatcovers at least a part of the feedback resistor. Also, an output of theguard amplifier may be connected to the shield.

In an equivalent circuit, the parasitic capacitance that couples withthe feedback resistor is connected to the signal line. With such anembodiment, by shielding the feedback resistor, and by supplying anelectric potential to the shield by means of the guard amplifier, suchan arrangement is capable of reducing the effects of the floatingcapacitance that couples with the feedback resistor.

Also, the shield may cover one electrode, which is arranged on an inputterminal side of the non-inverting amplifier, of the feedback resistorhaving two electrodes. Such an arrangement is capable of reducing thefloating capacitance that occurs between the shield and one terminal,which is arranged on the output terminal side of the invertingamplifier, of the feedback resistor having two electrodes.

Also, the current measurement circuit may further comprise a protectionelement arranged between the signal line and the shield line.

This also allows the effects of the parasitic capacitance that occursdue to the protection element to be reduced.

Also, the signal line and the shield line are configured as a cablehaving a hollow coaxial cable structure or otherwise a low-capacitancecoaxial cable structure. This allows the parasitic capacitance thatoccurs between the signal line and the shield line to be reduced.

Also, the current measurement circuit may be integrated on asemiconductor substrate together with a part of a device under test thatgenerates the current signal. Also, the signal line and the shield linemay be formed on the semiconductor substrate. This allows the parasiticcapacitance that occurs between the signal line and the shield line tobe reduced.

Also, the impedance circuit may further comprise a feedback capacitorarranged in parallel with the feedback resistor between an outputterminal of the inverting amplifier and the non-inverting input terminalof the non-inverting amplifier. Such an arrangement provides improvementof the stability of the system.

Another embodiment of the present invention relates to a base sequenceanalyzing apparatus. The base sequence analyzing apparatus comprises: anelectrode pair; a position control apparatus that controls a basesequence having a linear structure such that it passes through theelectrode pair; any one of the aforementioned current measurementcircuits that detect a current that flows between a gap of the electrodepair; and a data processing apparatus that analyzes an output of thecurrent measurement circuit.

Such an embodiment provides high-speed operation of the transimpedanceamplifier. Thus, such an arrangement allows the number of times eachbase is sampled to be increased, thereby providing improved measurementprecision. Alternatively, such an arrangement allows the number of basesthat can be analyzed per unit time to be increased.

It is to be noted that any arbitrary combination or rearrangement of theabove-described structural components and so forth is effective as andencompassed by the present embodiments.

Moreover, this summary of the invention does not necessarily describeall necessary features so that the invention may also be asub-combination of these described features.

BRIEF DESCRIPTION OF THE DRAWINGS

Embodiments will now be described, by way of example only, withreference to the accompanying drawings which are meant to be exemplary,not limiting, and wherein like elements are numbered alike in severalFigures, in which:

FIG. 1 is a circuit diagram showing a current measurement circuitincluding a transimpedance amplifier;

FIG. 2 is an equivalent circuit diagram showing the current measurementcircuit shown in FIG. 1;

Drawings:

FIG. 3 is a diagram showing the noise characteristics of thetransimpedance amplifier;

FIG. 4 is a circuit diagram showing a current measurement circuitaccording to a first embodiment;

FIGS. 5A through 5C are diagrams each showing a configuration of acable;

FIG. 6 is an equivalent circuit diagram showing the current measurementcircuit shown in FIG. 4;

FIG. 7A is a diagram showing the frequency characteristics of thecurrent measurement circuit shown in FIG. 4, and FIG. 7B is a diagramshowing the noise characteristics of the current measurement circuitshown in FIG. 4;

FIG. 8 is a circuit diagram showing a current measurement circuitaccording to a second embodiment;

FIG. 9A is a diagram showing the frequency characteristics of acorrection amplifier, FIG. 9B is a diagram showing the frequencycharacteristics of the current measurement circuit, and FIG. 9C is adiagram showing the noise characteristics of the current measurementcircuit;

FIG. 10 is a circuit diagram showing a current measurement circuitaccording to a third embodiment;

FIGS. 11A and 11B are diagrams each showing a part of a currentmeasurement circuit according to a first modification; and

FIG. 12 is a block diagram showing a base sequence analyzing apparatusincluding a current measurement circuit.

DETAILED DESCRIPTION OF THE INVENTION

The invention will now be described based on preferred embodiments whichdo not intend to limit the scope of the present invention but exemplifythe invention. All of the features and the combinations thereofdescribed in the embodiment are not necessarily essential to theinvention.

In the present specification, the state represented by the phrase “themember A is connected to the member B” includes a state in which themember A is indirectly connected to the member B via another member thatdoes not affect the electric connection therebetween, in addition to astate in which the member A is physically and directly connected to themember B.

Similarly, the state represented by the phrase “the member C is providedbetween the member A and the member B” includes a state in which themember A is indirectly connected to the member C, or the member B isindirectly connected to the member C via another member that does notaffect the electric connection therebetween, in addition to a state inwhich the member A is directly connected to the member C, or the memberB is directly connected to the member C.

First Embodiment

FIG. 4 is a circuit diagram showing a current measurement circuit 100according to a first embodiment. The current measurement circuit 100converts a current signal I_(IN) output from a DUT 200 into a voltagesignal V_(OUT). The current measurement circuit 100 includes a cable110, a non-inverting amplifier 120, an inverting amplifier 130, animpedance circuit 140, and a guard amplifier 150.

A signal line 112 is configured as a path via which the current signalI_(IN) is transmitted. The signal line 112 is arranged such that one endthereof is connected to a DUT 200 and the other end thereof is connectedto an input terminal I1 of the non-inverting amplifier 120. A shieldline 114 is arranged in the vicinity of at least a part of the signalline 112. At least a part of the signal line 112 and at least a part ofthe shield line 114 may be housed in the same cable 110. It should benoted that the state represented by “the signal line and the shield lineare arranged in the vicinity of each other” represents a state in whichthe signal line and the shield line are arranged within a range ofdistance in which a parasitic capacitance that occurs between the signalline and the shield line has a significant effect on the gain and thefrequency characteristics of the current measurement circuit.Specifically, such a distance is on the order of several hundreds of μmto several hundreds of mm.

FIGS. 5A through 5C are diagrams each showing a configuration of thecable 110. For example, the cable 110 may be configured as a hollowcoaxial cable as shown in FIG. 5A. Also, the cable 110 may be configuredas a hollow triaxial cable as shown in FIG. 5B. Also, the cable 110 maybe configured as a low-capacitance coaxial cable as shown in FIG. 5C.With the hollow coaxial cable shown in FIG. 5A or 5B, the cable coreconfigured as the signal line 112 is supported by a support member 116formed of a Teflon (trademark) material or the like with respect to anelectrically conducting pipe configured as the shield line 114. Byemploying such a cable, such an arrangement allows the absolute value ofthe parasitic capacitance that occurs between the signal line 112 andthe shield line 114 to be reduced.

Returning to FIG. 4, the non-inverting amplifier 120 amplifies theelectric potential at its input terminal I1 without inversion, andoutputs the output signal via its output terminal O1. The non-invertingamplifier 120 includes an operational amplifier 122. The operationalamplifier 122 is arranged such that its non-inverting input terminal (+)is connected to the signal line 112, and its inverting input terminal(−) receives its output signal O1 as a feedback signal. Morespecifically, an output signal obtained by dividing the output signal O1by means of resistors R1 and R2 is fed back to the inverting inputterminal of the operational amplifier 122. The gain of the non-invertingamplifier 120 is represented by (R1+R2)/R2. Also, the non-invertingamplifier 120 may be configured to have a gain on the order of 100 (40dB).

The inverting amplifier 130 amplifies the output signal O1 of thenon-inverting amplifier 120 with inversion so as to generate an outputvoltage V_(OUT), and outputs the generated output voltage V_(OUT) viaits output terminal O2. The open loop gains A_(OL) of the non-invertingamplifier 120 and the inverting amplifier 130 are preferably set tovalues on the order of 60 to 80 dB. The non-inverting amplifier 120 andthe inverting amplifier 130 are each configured to have an appropriategain.

The impedance circuit 140 includes a feedback resistor R_(F) arrangedbetween the output terminal O2 of the inverting amplifier 130 and theinput terminal I1 of the non-inverting amplifier (non-inverting inputterminal of the operational amplifier 122). Furthermore, the feedbackcapacitor C_(F) is arranged in parallel with the feedback resistorR_(F).

The guard amplifier 150 receives the electric potential V− at theinverting input terminal (−) of the operational amplifier 122, andstabilizes the electric potential at the shield line 114 to be the sameas the electric potential V− at the inverting input terminal of theoperational amplifier 122. The guard amplifier 150 may be configured asa voltage follower. Also, the guard amplifier 150 may have otherconfigurations.

The above is the configuration of the current measurement circuit 100.FIG. 6 is an equivalent circuit diagram showing the current measurementcircuit 100 shown in FIG. 4. The advantages of the current measurementcircuit 100 can be clearly understood in comparison with the equivalentcircuit diagram in FIG. 6 and the equivalent circuit diagram in FIG. 2.It should be noted that a circuit block 800A shown in FIG. 6 includingthe non-inverting amplifier 120, the inverting amplifier 130, and theimpedance circuit 140 corresponds to the transimpedance amplifier 800shown in FIG. 2. As further seen in FIG. 6, the DUT 200 can be modeledas a circuit comprising a current source 202 that generates a tunnelcurrent i_(DUT), a parasitic parallel resistor R_(DUT), and a parasiticparallel capacitor C_(DUT). Additionally, the operational amplifier 122of the non-inverting amplifier 120 included common input capacitancesCMN1 and CMP1 at its input.

The current measurement circuit 100 has a two-stage configurationcomprising the non-inverting amplifier 120 and the inverting amplifier130. In FIG. 2, the differential input capacitance CD of the operationalamplifier 802 is connected in parallel with the cable capacitanceC_(CAB) or the like, which functions as a part of the input shuntcapacitance C_(S). This leads to a reduction in the cutoff frequency f2of the transimpedance amplifier 800. In contrast, with the equivalentcircuit shown in FIG. 6, the differential input capacitance CD1 of theoperational amplifier 122 is not arranged in parallel with the cablecapacitance C_(CAB) or the like. Thus, such an arrangement is capable ofpreventing the cutoff frequency f2 of the transimpedance amplifier 800Afrom degrading due to the differential input capacitance CD1.

Furthermore, by arranging the shield line 114 in the vicinity of thesignal line 112, such an arrangement is capable of reducing thecapacitance that occurs between the signal line 112 and the ground. Thisallows the input shunt capacitance C_(S) to be reduced.

Furthermore, by virtually grounding the operational amplifier 122, theelectric potential at the inverting input terminal (−) of theoperational amplifier 122 becomes substantially the same as that at thenon-inverting input terminal (+) thereof. With such an arrangement, thesignal line 112 is connected to the non-inverting input terminal (+) ofthe operational amplifier 122. Furthermore, the electric potential atthe inverting input terminal (−) of the operational amplifier 122 isapplied to the shield line 114 via the guard amplifier 150. Thus, theelectric potential at the signal line 112 becomes substantially the sameas that at the shield line 114. With such an arrangement, the voltagedifference between the signal line 112 and the shield line 114 ismaintained at a given value (substantially at 0 V) even if theyfluctuate. Thus, the parasitic capacitance C_(CAB) is neither chargednor discharged. Thus, in the equivalent circuit, the parasiticcapacitance C_(CAB) becomes zero, thereby reducing the effect of theparasitic capacitance C_(CAB).

Furthermore, with the present embodiment, the input terminal of theguard amplifier 150 is connected to the inverting input terminal (−) ofthe operational amplifier 122 of the non-inverting amplifier 120,instead of being connected to the non-inverting input terminal (+)thereof. With such an arrangement, the input capacitances CMN2, CMP2,and CD2 of the guard amplifier 150 are not coupled with the signal line112. Thus, such input capacitances do not lead to an increase in thecapacitance C_(S) as viewed from the signal line 112. That is to say, itcan be said that the guard amplifier 150 has substantially no effect onthe cutoff frequency f2.

FIG. 7A is a diagram showing the frequency characteristics of thecurrent measurement circuit 100 shown in FIG. 4. FIG. 7B is a diagramshowing the noise characteristics of the current measurement circuit 100shown in FIG. 4. The frequency characteristics and the noisecharacteristics were calculated by simulation.

As shown in FIG. 7B, the current measurement circuit 100 is capable ofgreatly reducing the effect of the parasitic capacitance C_(S) as viewedfrom the signal line 112, as compared with an arrangement shown inFIG. 1. This allows the noise gain to be reduced.

As a comparison technique, another two-stage configuration isconceivable in which the first stage is configured as a voltage followerhaving a gain of 1, and the second stage is configured as a high-gainamplifier. However, it is difficult for the second-stage amplifier tosecure a sufficient open loop gain A_(OL) over a wide bandwidth.Furthermore, such a configuration leads to a problem of increased noise,and leads to difficulty in maintaining stable operation of the system.With the current measurement circuit 100 according to the embodiment,the transimpedance amplifier 800A has a two-stage configurationcomprising the non-inverting amplifier 120 and the inverting amplifier130. Furthermore, by configuring the first-stage amplifier, i.e., thenon-inverting amplifier 120, to have a high gain, such an arrangementprovides the transimpedance amplifier 800A with a high gain over a widebandwidth while suppressing an increase in noise.

Second Embodiment

FIG. 8 is a circuit diagram showing a current measurement circuit 100 aaccording to a second embodiment. The current measurement circuit 100 afurther includes a protection element 160 and a correction amplifier 170in addition to the components included in the current measurementcircuit 100 shown in FIG. 4. Typically, such a protection element 160 isarranged between the signal line 112 and the ground line. However, inthe present embodiment, the protection element 160 is arranged betweenthe signal line 112 and the shield line 114.

With such an arrangement, as with the parasitic capacitance C_(CAB) thatoccurs due to the cable 110, a parasitic capacitance C_(PRO) that occursdue to the protection element 160 becomes substantially zero as viewedfrom the signal line 112. That is to say, such an arrangement allows theeffects of the parasitic capacitance C_(PRO) to be reduced. Thus, suchan arrangement is capable of preventing an increase in the capacitanceC_(S) connected to the signal line 112 even in a case in which such aprotection element 160 is provided.

The correction amplifier 170 corrects the frequency characteristics ofthe output signal O2 output from the inverting amplifier 130, andoutputs an output voltage V_(OUT). FIG. 9A is a diagram showing thefrequency characteristics of the correction amplifier 170. FIG. 9B is adiagram showing the frequency characteristics of the current measurementcircuit 100 a of FIG. 8. FIG. 9C is a diagram showing the noisecharacteristics of the current measurement circuit 100 a of FIG. 8. Aspecific configuration of the correction amplifier 170 is not restrictedin particular. Such a correction amplifier 170 may preferably beconfigured as a known high-emphasis filter or the like.

As shown in FIG. 7B, the transimpedance amplifier 800A has a two-stageconfiguration comprising the non-inverting amplifier 120 and theinverting amplifier 130 so as to reduce the capacitance C. This reducesthe noise gain of the transimpedance amplifier 800A in a high-frequencyrange. Thus, by boosting the high-frequency component by means of thecorrection amplifier 170, such an arrangement provides a sufficientlyhigh signal gain over a wide bandwidth while maintaining the noise gainat the same level as that provided by conventional arrangements. FIG. 9Ashows a case in which the bandwidth is widened from a value on the orderof 10 kHz to 100 kHz.

It should be noted that the protection element 160 or otherwise thecorrection amplifier 170 may be omitted from the current measurementcircuit 100 a shown in FIG. 8. Also, such an arrangement is effective asan embodiment of the present invention.

Third Embodiment

FIG. 10 is a circuit diagram showing a current measurement circuit 100 baccording to a third embodiment. The current measurement circuit 100 bfurther includes a shield 180 in addition to the components of thecurrent measurement circuit 100 a shown in FIG. 8. The feedback resistorR_(F) is configured as a chip element or otherwise an SIP (Single InlinePackage) element. The feedback resistor R_(F) includes a pair of exposedelectrodes E1 and E2. The shield 180 is arranged such that it covers atleast a part of the feedback resistor R_(F). The shield 180 ispreferably arranged such that it covers the electrode E1 of the feedbackresistor R_(F) on the input side of the non-inverting amplifier 120. Theoutput of the guard amplifier 150 is electrically connected to theshield 180. The electric potential at the inverting input terminal ofthe operational amplifier 122 is applied to the shield 180.

In an equivalent circuit of the current measurement circuit 100 b, thecoupling of the floating capacitances Cx and Cy to the feedback resistorR_(F) is connected to the signal line 112. With the current measurementcircuit 100 b, by shielding the feedback resistor R_(F), and bysupplying the same electric potential as that at the signal line 112 tothe shield by means of the guard amplifier 150, such an arrangement iscapable of reducing the effects of the coupling of the floatingcapacitances Cx and Cy to the feedback resistor R_(F).

Furthermore, by configuring the shield 180 such that it covers only theelectrode E1 on the input terminal side of the non-inverting amplifier120, such an arrangement allows the floating capacitance Cy that occursdue to the electrode E2 on the output terminal side to be reduced. Suchan arrangement is capable of preventing the bandwidth from narrowing.

Description has been made regarding the present invention with referenceto the embodiment. The above-described embodiment has been described forexemplary purposes only, and is by no means intended to be interpretedrestrictively. Rather, it can be readily conceived by those skilled inthis art that various modifications may be made by making variouscombinations of the aforementioned components or processes, which arealso encompassed in the technical scope of the present invention.Description will be made below regarding such modifications.

First Modification

Description has been made in the embodiments regarding an arrangement inwhich a part of the signal line 112 and a part of the shield line 114are housed in the cable 110. However, the present invention is notrestricted to such an arrangement. FIGS. 11A and 11B are diagrams eachshowing a part of a current measurement circuit 100 c according to afirst modification.

In this modification, the current measurement circuit 100 c isintegrated on a single semiconductor substrate 210 together with a partof the device under test 200 that generates the current signal I_(IN).Such an arrangement does not require the cable 110. The signal line 112and the shield line 114 are each formed as an LSI line on thesemiconductor substrate 210. FIG. 11A is a plan view of the currentmeasurement circuit 100 c. The shield line 114 may be formed adjacent tothe signal line 112. More preferably, the shield lines 114 may be formedsuch that the signal line 112 is interposed between them. FIG. 11B is across-sectional view of the current measurement circuit 100 c. Also,such a shield line 114 may be formed in a wiring layer adjacent to thatincluding the signal line 112 such that they overlap. Also, such ashield line 114 may be formed in an upper wiring layer adjacent to thatincluding the signal line 112.

Such a modification allows the parasitic capacitance C_(CAB) that occursbetween the signal line 112 and the shield line 114 to be reduced, ascompared with an arrangement employing the cable 110.

Second Modification

The configuration of the impedance circuit 140 is not restricted to suchan arrangement as described in the embodiments. Also, various kinds ofother configurations may be employed.

[Usage]

Lastly, description will be made regarding the usage of the currentmeasurement circuit 100. The current measurement circuit 100 accordingto the embodiment may be employed in a base sequence analyzing apparatus(DNA sequencer or RNA sequencer) 300. FIG. 12 is a block diagram showingthe base sequence analyzing apparatus 300 including the currentmeasurement circuit 100. The base sequence analyzing apparatus 300includes an electrode pair 310, a position control apparatus 320, theaforementioned current measurement circuit 100, and a data processingapparatus 330.

The DUT 200 is configured as a chip including the electrode pair 310, ananopore structure 312, an electrophoresis electrode pair 314, anunshown nanochannel, an unshown nanopillar structure, and the like. ADNA sample is controlled such that it passes through the nanochannel, soas to separate and extract a single DNA molecule. After the DNA moleculepasses through the nanopillar structure, the DNA molecule can beanalyzed as a linear sample.

The electrophoresis electrode pair 314 and a driving amplifier 316 formthe position control apparatus 320 that controls the position of the DNAmolecule 204. The position control apparatus 320 applies an electricfield to the DNA molecule so as to move the DNA molecule such that itpasses through a gap between the electrode pair 310 formed in thenanopore structure 312.

When a base of the DNA molecule passes through the gap between theelectrode pair 310, the tunnel current I_(IN) flows according to thebase type. The current measurement circuit 100 detects the tunnelcurrent (current signal) I_(IN), and converts the current signal thusdetected into the voltage signal V_(OUT). The voltage signal V_(OUT) isconverted by an A/D converter 318 into a digital value. The digitalvalue thus converted is input to the data processing apparatus 330. Thedata processing apparatus 330 is configured as a computer includingmemory and a processor. The data processing apparatus 330 performssignal processing so as to identify the base sequence of the DNAmolecule.

As described above, the current measurement circuit 100 according to theembodiment provides low-noise characteristics and high-speed operation.Thus, such an arrangement allows the number of times each base issampled to be increased, thereby providing improved measurementprecision. Alternatively, such an arrangement allows the number of basesthat can be analyzed per unit time to be increased. Also, such anarrangement is capable of reducing noise that occurs in the currentmeasurement circuit 100. Thus, such an arrangement allows the requirednumber of times each base is sampled to be reduced.

Description has been made with reference to FIG. 12 regarding asequencer using the gating nanopore method. Also, the currentmeasurement circuit 100 may be applied to a sequencer using the MCBJmethod.

While the preferred embodiments of the present invention have beendescribed using specific terms, such description is for illustrativepurposes only, and it is to be understood that changes and variationsmay be made without departing from the spirit or scope of the appendedclaims.

What is claimed is:
 1. A current measurement circuit that converts acurrent signal into a voltage signal, the current measurement circuitcomprising: a signal line via which the current signal is transmitted; ashield line arranged in the vicinity of at least a part of the signalline; a non-inverting amplifier comprising an operational amplifierhaving its non-inverting input terminal supplied with the current signalthe signal line and its inverting input terminal supplied with itsoutput signal as a feedback signal; an inverting amplifier thatamplifies the output signal of the non-inverting amplifier withinversion so as to generate the voltage signal; an impedance circuitcomprising a feedback resistor arranged between an output terminal ofthe inverting amplifier and the non-inverting input terminal of theoperational amplifier; and a guard amplifier that receives an electricpotential at the inverting input terminal of the operational amplifier,and that applies the electric potential to the shield line.
 2. Thecurrent measurement circuit according to claim 1, further comprising acorrection amplifier that corrects frequency characteristics of anoutput signal of the inverting amplifier.
 3. The current measurementcircuit according to claim 1, further comprising a shield that covers atleast a part of the feedback resistor, wherein an output of the guardamplifier is connected to the shield.
 4. The current measurement circuitaccording to claim 3, wherein the shield covers one electrode, which isarranged on an input terminal side of the non-inverting amplifier, ofthe feedback resistor having two electrodes.
 5. The current measurementcircuit according to claim 1, further comprising a protection elementarranged between the signal line and the shield line.
 6. The currentmeasurement circuit according to claim 1, wherein the signal line andthe shield line are configured as a cable having a hollow coaxial cablestructure or otherwise a low-capacitance coaxial cable structure.
 7. Thecurrent measurement circuit according to claim 1, wherein the currentmeasurement circuit is integrated on a semiconductor substrate togetherwith a part of a device under test that generates the current signal,and wherein the signal line and the shield line are formed on thesemiconductor substrate.
 8. The current measurement circuit according toclaim 1, wherein the impedance circuit further comprises a feedbackcapacitor arranged in parallel with the feedback resistor between anoutput terminal of the inverting amplifier and the non-inverting inputterminal of the non-inverting amplifier.
 9. A base sequence analyzingapparatus comprising: an electrode pair; a position control apparatusthat controls a base sequence having a linear structure such that itpasses through the electrode pair; the current measurement circuitaccording to claim 1, that detects a current that flows between a gap ofthe electrode pair; and a data processing apparatus that analyzes anoutput of the current measurement circuit.